This invention relates to the field of electronic switching. In particular, the invention provides an improvement in a pulse-width modulated circuit for driving a load, such as an electric motor.
The prior art includes many examples of pulse-width modulation (PWM) as a means of controlling the current through a load. For example, U.S. Pat. Nos. 5,070,292 and 5,081,409 describe such PWM circuits. This specification incorporates by reference the disclosures of the latter patents. In these patents, and in similar PWM circuits of the prior art, a stream of pulses controls electronic switches which open and close different circuit paths for applying current to the load. The widths of the pulses determine when, and in what direction, the circuit applies current to the load. Thus, the pulse widths directly control the effective current in the load.
This invention addresses the problem caused by the fact that electronic switches and diodes do not change states instantaneously. Most theoretical discussions of PWM ignore the transient behavior of electronic switches, but these transient effects cause losses and reduce the efficiency of the PWM system.
Switching losses comprise power losses incurred by the electronic switch as it turns on or off. The power loss becomes dissipated in the switch in the form of heat. The present invention has as its major object the reduction of such switching losses.
The "reverse recovery" effect forms a significant component of switching losses in an electronic switch. A semiconductor diode presents a low electrical resistance to current flowing in one direction (the "forward direction") and a high resistance to current flowing in the opposite direction (the "reverse direction"). One intends that current should flow substantially unimpeded in the forward direction, and that virtually no current should flow in the reverse direction. Although the latter statements hold true for the steady-state condition, these statements do not describe what happens in the very short term. It turns out that when one forward-biases the diode and then changes the applied voltage so as to reverse-bias the diode, current through the diode does not immediately cease. In fact, for a short time after the biasing has reversed, current actually flows in the opposite direction, and with little resistance. The magnitude of this reverse current approximately equals the magnitude of the current that flowed in the forward direction, as the reverse current momentarily "sees" virtually no resistance. As time passes, however, the reverse current rapidly diminishes, until the reverse-biased diode reaches a state wherein virtually no current passes through it. This reverse current diminishes in magnitude very quickly, and normally has little effect on the operation of an electronic circuit. However, in applications where one must perform rapid and repeated switching, the effect can have serious consequences. The reverse recovery effect varies according to the design of the diode; some diode designs can substantially reduce the amount of reverse current. In general, one observes the reverse recovery effect when one switches a diode from forward biasing to reverse biasing.
Another practical problem arises from the inherent properties of semiconductor switching devices such as field-effect transistors (FETs). The source and drain of a FET together inherently function as a diode and a FET in parallel. One can draw the equivalent circuit for a MOSFET as a MOSFET connected in parallel with a diode. One calls this unintended diode a "parasitic diode", because the diode forms part of every FET, due to the inherent construction of the FET. Unfortunately, such parasitic diodes normally have very poor reverse recovery effects. By "poor" one means that the parasitic diode of a FET passes a reverse current having a large magnitude and/or allows that current to flow for a relatively long time, and that the current does not turn off smoothly when it finally does turn off. The latter behavior of a diode is analogous to the phenomenon of "water bang" observed in a household plumbing system. When a water faucet is shut off rapidly, the entire column of water constrained to the pipe must stop abruptly, causing the pipes to jerk suddenly to dissipate the energy stored in the moving water. The latter effect may eventually result in damage to the plumbing system. Analogous stresses develop in the above-described electrical circuit due to the sudden commutation of the internal diode.
Thus, the relatively poor internal diodes reduce the efficiency of a high-speed switching circuit. This is because the unwanted current flows while the switch is operated in the "linear region" wherein voltage is still dropping across it while current is flowing, resulting in high power dissipation.
FIG. 1 illustrates a typical circuit of the prior art. This circuit constitutes an "H-bridge", formed of four switches M1, M2, M3, and M4. Each switch includes a FET, as well as two diodes, one in series with the FET and the other in parallel with the combination of the FET and the series diode. The FETs also explicitly show the parasitic diodes inherent in each FET. These parasitic diodes do not comprise separate components, but appear in the figure for emphasis. Diodes D2, D4, D6, and D8 provide alternative paths for current instead of through the parasitic diodes; in practice, one chooses D2, D4, D6, and D8 to have better turn on and/or turn off characteristics than those of the parasitic diodes. Note that the reverse recovery characteristic forms part of the turn off characteristic of the diode.
A power supply (not shown) supplies voltage +V, and the current flows back to the power supply through the ground connection shown.
One applies a PWM signal, shown in the pulse diagram of FIG. 2, to the switches M1, M2, M3, and M4, shown in FIG. 1. In particular, the basic PWM signal x controls switches M1 and M4, while the complemented PWM signal (x') controls switches M2 and M3. In practice, signal x' does not exactly represent the complement of PWM; instead, x' becomes high shortly after x becomes low, and x' becomes low shortly before x becomes high. The latter arrangement prevents temporary short circuits across the power supply, which would occur, for example, with both M1 and M2, or M3 and M4, in the conductive state. FIG. 2 shows this "dead time" between x and x' pulses in an exaggerated manner.
Now assume that current flows through the load from left to right; one can call this direction the positive direction. Suppose that x has reached its high state. Then M1 and M4 conduct, and current flows from the power supply (+V), through M1 and D1, through the load, through M4 and D7, and back to the power supply via ground. At this moment, the voltage at point A approximately equals +V, and the voltage at B approximately equals ground. Thus the voltage applied across the load (V.sub.A -V.sub.B) equals about +V volts. Due to the inductive nature of the load, we can say that (di/dt): V/L, where i represents the current through the load and L represents the inductance of the load. Thus, the current increases in magnitude, in the positive direction, at the approximate rate of V/L.
Next, consider what happens when x first becomes low. At this moment, x' initially remains low, as explained above. For an inductive load, current continues to flow through the load, as it cannot change instantaneously. For very short time periods, one can consider the load as a constant current source, due to the tendency of the inductor to resist change in its magnetic field. M1 and M4 take a finite amount of time to turn off. As M1 and M4 turn off, they become more resistive as time passes. Since the current remains nearly constant during this short time period, the more resistive M1 and M4 become, the greater the voltage drop across M1 and M4, until the voltage at point A becomes less than ground and the voltage at point B becomes greater than +V. At this time, D4 and D6 become forward-biased and the load current now flows from ground, through D4, through the load, through D6, and to the power supply (+V). Current will not flow through M2 or M3 because diodes D5 and D3 block current flow from source to drain through the FETs.
The magnitude of the current flowing through the load decreases because one has effectively applied the negative of the voltage formerly applied to the load (note that point A represents ground potential when D4 conducts, so the power supply voltage +V appears at the side of the load near point B). The rate at which the current decreases approximately equals (di/dt)=(V.sub.A -V.sub.B)/L=-(+V)/L.
When x' becomes high, the FETs associated with M2 and M3 conduct, but due to D3 and D5, current continues to flow in the direction indicated above, through D4, the load, and D6. Similarly, when x' becomes low, current continues to flow in the same manner.
Next, as x becomes high again, the FETs associated with M1 and M4 begin to turn on. As they turn on, they become less resistive. In becoming less resistive, they begin to conduct more and more current. They will continue to conduct increasing amounts of load current until the switches carry the full load current. At this time, D4 and D6 become reverse-biased. The power supply now becomes momentarily short-circuited, because current can flow through M1 and D1, and then through D4 (during the reverse recovery time for this diode). A similar short circuit occurs through D6, M4, and D7, because of the reverse recovery current of D6. Note that M1 and M4 carry the respective reverse recovery currents (of D4 and D6, respectively) plus the load current. As the reverse recovery currents in D4 and D6 decrease to zero, the current flowing through M1 and M4 becomes the load current only.
One can explain the switching losses in the H-bridge circuit of FIG. 1 in the following manner. When x goes low, M1 and M4 begin to turn off, and as they turn off they become more resistive. At the same time, the current through M1 and M4 does not change. The latter facts translate into excess power losses. The power loss equals the voltage across the device turning off multiplied by the current flowing through the device.
Similarly, one experiences a switching loss when M1 and M4 turn on. As M1 and M4 turn on, they become less resistive. In becoming less resistive, they begin to draw increased amounts of current. They will continue to draw increasing amounts of current until the switch conducts the full load current. At this time, D4 stops conducting and becomes reverse-biased, causing a reverse recovery current spike. This current spike flows through M1 and M4 in addition to the load current. The switching power loss in the FETs equal the voltage across the device turning off multiplied by the current flow through the device.
A similar analysis applies in the case where current flows through the load in the negative direction (from point B to point A). In this case, the dominant current path (which occurs when M2 and M3 conduct, when x' becomes high) comprises M3, D5, the load, M2, and D3, and the path of current when all FETs cease to conduct (and also when M1 and M4 are turned on) would include D8, the load, and D2. Similar switching losses occur when the switches change state, for the same reasons given in the case of positive current flow.
The invention disclosed in U.S. patent application Ser. No. 08/015,531, cited above, reduces the switching losses in a PWM switching device. The disclosure of the latter application is hereby incorporated by reference. The present invention provides further improvements over the circuit disclosed in application Ser. No. 08/015,531.